Tiny low power flyback converters
06 August 2008
When challenged by the requirements for high input voltage capability, only a flyback solution using a simple controller and an off-the-shelf transformer will do
The designer of a DC/DC converter in industrial, medical and automotive applications is always going to be
challenged by the requirement for an input voltage capability up to 42V.
Often the output power of a system is quite low (<1W) so under normal circumstances, a buck converters could handle the power; especially as some are capable of handling high voltages.
They are easy to use and come in small sizes, but do not isolate the input. The recommendation in this situation would be to use a flyback topology, which provides the necessary isolation and flexible power
handling. However, a drawback is the complex design of the transformer and its customisation.
Therefore, a flyback solution using a simple controller and a standard off-the-shelf transformer would seem to be the answer. As cost is always a concern, this is seen as the optimum choice. Due to the low power,
the designer is also challenged to make the application as small as possible, similar to non-isolated topologies. A greater challenge for a small PCB area is the requirement for an operating temperature between 85°C and 105°C.
Power supply requirements In factory automation the main requirements of a power supply are:
Vin nom operating: 24V
Vin operating: 18V - 30.5V
Vin transient: 36V - 42V
T ambient operating: -40°C to +85°C
The challenge for the designer is to protect the system from transients; usually achieved by employing transient suppresser devices. As their specification for voltage clamping is not exact, the best system
protection is to find ICs that can withstand the above mentioned transients. In automotive the main requirements of a power supply are:
Vin nom operating: 12V
Vin operating: 6V - 18V
Vin cold crank: 3V
Vin transient: 36V - 42V
T ambient operating: -40°C to +105°C
The challenge here is even tougher because the application not only has to survive the transients, but also work during cold cranking. This is becoming increasingly important due to the growing numbers of cars with ‘stop-start’ systems that turn off the engine when the vehicle is stationary and re-start it when required.
In medical systems the 24V bus is often used to distribute power to the point of load. The voltage requirements are therefore equivalent to the factory automation ones.
Application design
The specifications of the design are:
Input: 6 to 40VDC, 24VAC
Output: 3.3V@100mA
Power: 330mW
Switching frequency: 250kHz
Isolation: Input/output: 500V
Ambient temperature: -40°C to +105°C
Maximum component height: 11mm
Overall size: 20mm x 36.5mm
At the heart of the flyback power supply is the LM3481 current mode controller from National Semiconductor (see figure 1).
The bridge rectifier is needed in case of AC input. The voltage transient suppresser clamps high input voltages and protects the entire system. The design uses a standard transformer that provides energy to the
output and an optocoupler to feed back the error of the output voltage to the primary side controller.
Inrush behaviour
At plug-in, the input voltage charges the input capacitor with a low peak current due to the DCR of the input filter. The mains filter is located at the input of the converter consisting of L1, L2, C1 and is necessary to
minimise interference from the power supply to the network and vice versa.
In case of multiple point of loads in a network, the newly connected power supply must wait a minimum time before starting the DC/DC converter. This is ensured by a divider connected to the UVLO pin (under
voltage lock out) of the LM3481 and a capacitor C15 that is charged by Vin through R1. As soon as the voltage across the capacitor reaches the UVLO threshold, the flyback starts switching with narrow pulses
and gradually becomes wider. As the duty cycle increases, the output voltage also increases and linearly charges up the output capacitor.
For automotive applications in cold crank conditions, the minimum input voltage of 3V can be achieved due to the controller’s capability to work down to this level. Decreasing the sense resistance should increase the current sense limit.
Switching cycle description
Figure 3 shows the drain source voltage and drain current of Q2 for one complete cycle in ON mode and illustrates the different phases.
Switch ON phase (1) is when the internal driving circuit switches on the power MOSFET Q2 and the voltage across it is almost zero. The parasitic capacitor of the MOSFET is discharged, which explains the large drain current peak at the start of the conduction phase and a source of power losses. Conducting phase (2) is where during TON the current rises linearly until reaching the value defined by IL = (VIN/Lp)*TON. Switch off phase (3) is when the MOSFET switches off and the current is abruptly cut off. At this time there is a large current flowing through the primary winding. This causes the voltage across the MOSFET to rise very
quickly. The clamp circuit defined by D5 and D6 limits the maximum voltage across the mosfet. The off
phase (4) is divided into two parts; the energy transfer phase and the dead time. On phase (5), the secondary diode conducts and the magnetic energy that is stored in the transformer is transmitted to the output capacitor with linear falling current of D8.
During this energy transfer phase, the voltage at the drain is equal to the input voltage across the input
capacitor plus the secondary voltage multiplied by the turn ratio of the transformer.
After the energy transfer is complete, there is a dead time (6), which continues until the next conduction
pulse occurs. During this time, the MOSFET and the secondary diodes are off. The voltages are oscillating
because of primary winding inductance and the MOSFET drainsource capacitor. Snubbers are used on the primary and secondary sides to keep EMI within required limits.
The cycle by cycle current limitation on the primary side controls the maximum transferable power. The secondary voltage will drop down if this limit is reached. The voltage across resistor R8 is proportional to the drain current and is sensed by the input pin 1 of the LM3481. If the voltage at R8 becomes high enough, the PWM of U1 uses this information to terminate the mosfet conduction. To protect the power supply input, a varistor RV1 is included to absorb any high peak voltages.
During a short circuit the power supply provides the maximum current defined approximately by the fold back point. As soon as the short circuit is removed from the output, the power supply will return to its
regulated output voltage. If there is a failure, such as an open loop of the PWM regulation, the 3.3V increases and the voltage comp pin 3 becomes higher and switches off the MOSFET. As the internal over voltage protection is disabled due to the pin VFB connected to ground, the output voltage wil increase, reaching 6.3V at 100mA. To ensure the reliability and life time, a 20 per cent derating (current, voltage, temperature) has been defined for all components.
Having calculated the customised transformer values using software tools, a standard flexible transformer (six identical windings) was selected. The windings were then combined on the schematic in such a way that the total inductance, peak current and turn ratio matches the customised values. The regulation loop is designed to have a cross over frequency of 2.5KHz and a phase margin of 90°, which ensures system
stability.
These parameters are set by choosing the appropriate values of the voltage divider R12 and R13, and the compensation network R5 C5 C8 R9 C9. R11 set the current on the LMV431 precision shunt regulator. The
optocoupler is needed to ensure feedback of the output voltage to the controller and to keep the isolation between the primary and the secondary side.
RALF REGENHOLD is principal product applications engineer, Europe, National Semiconductor. KAMAL NAJMI is power design engineer, power application design centre, Europe, National Semiconductor
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